With increased demand for faster response and reduced size and weight, switching frequency of dc-dc converters is increasing toward the megahertz range. The switching frequency of PWM converters was able to increase along with the speed of available switching semiconductor devices so long as the loss due to voltage/current overlap was the dominant limit on switching frequency. While faster power transistors have mitigated overlap loss, the reverse-recovery problem of power diodes as well as voltage spikes and discharging of the parasitic capacitances start to be limiting factors.
The buck converter shown in FIG. 1(a) is operated in a continuous conduction mode (CCM) and typical waveforms are shown as an example in FIG. 1(b). Like in any other PWM converter, semiconductor switching devices exhibit capacitive turn-on and inductive turn-off switching. Reverse-recovery current of the diode, discharging current of the parasitic capacitance across the transistor and energy stored in parasitic inductance do not contribute to power transfer but cause additional loss and produce voltage spikes and high-frequency parasitic oscillations as illustrated by waveforms in FIG. 2.
The problem of voltage spike and discharging parasitic capacitance are both consequences of the discontinuity of the waveforms of PWM converter as shown in FIGS. 1(b) and FIG. 2. As long as the converter has discontinuous waveforms, these two effects severely limit the performance at high switching frequencies. Only if the current and voltage are smooth and slowly varying the parasitic inductance and parasitic capacitance can be charged and discharged smoothly without losses.
One prior-art buck converter uses a noncontrollable saturable reactor in series with a rectifier diode as a magnetic snubber ("spike killer") to reduce reverse recovery current and its undesirable effects, O. Arakawa, T. Yamada and R. Hiramatsu, "Magnetic Snubber using Amorphous Saturable Reactor--Amorphous Beads," IEEE Applied Power Electronics Conference, (IEEE Publication 88CH2504-9), pp. 334-340, 1988 Record. Due to the nonlinear, square-loop characteristic of the magnetic core, a saturable reactor acts as a short circuit when saturated, thus not affecting power transfer. During the diode's turn-off, small negative diode current runs the saturable reactor out of saturation which represents large inductance and limits the slope of the reverse diode current di/dt, as well as its magnitude I.sub.DRM. By this action, turn-off losses in the diode and consequently turn-on loss of the transistor due to the diode's reverse recovery current, as well as parasitic oscillations and voltage spike on the diode, are significantly reduced.
In order to solve problems associated with parasitic reactance, an objective of this invention is to prevent voltage and current waveforms from being discontinuous, i.e., to make the converter insensitive to these parasitic components, namely inductances and capacitances. Different converter topologies have been proposed as a solution to this problem in prior-art converters, such as resonant converter topologies and soft-switching converter topologies.
A resonant converter is a power converter in which one or more switching waveforms (either voltage or current) contain pieces of sinusoidal ringing waveforms. It means that a switching waveform is a continuous, large ripple voltage or current that starts from zero and returns to zero. Due to this, switching loss caused by the voltage/current overlap as well as due to parasitic reactances can be effectively eliminated so the switching frequency can be increased.
A large number of various resonant converter topologies and their derivatives, such as quasi-resonant and multiresonant converters, have been proposed in the prior art. Even though these topologies are superior in reducing switching losses, they all suffer from very serious drawbacks, such as: (a) large circulating reactive energy, (b) increased component stresses, (c) increased conduction loss, (d) restricted operating region, (e) difficult analysis, (f) variable switching frequency, and (g) complex control.
Considering the advantages and disadvantages for both PWM and resonant converter topologies, the "ideal" topology would combine the best features of both topologies. This includes low switching losses, constant frequency operation, low component stresses, reasonable rated reactive components, low level of generated EMI noise, and a wide control and load range. Several families of switching converters that combine PWM and resonant behavior are classified as soft-switching converters.
Resonant elements in soft-switching converters are not used for energy transfer, but rather for shaping the current or voltage during switching transitions in order to provide either zero-current or zero-voltage switching. During the rest of the switching period, the converter behaves like a PWM converter. The resonant frequency is well above the switching frequency, and the resonant voltage is "clamped" by a rectifier or second switch after a switching transition is completed. The waveforms between transitions look very much like those in PWM converters.
One of the prior-art methods which provides soft-switching in basic dc-to-dc converter topologies is explained in C. P. Henze, H. C. Martin and D. W. Parsley, "Zero-Voltage Switching in High-Frequency Power Converters Using Pulse-Width Modulation," IEEE Applied Power Electronics Conference, (IEEE Publication 88CH2504-9), pp. 33-40, 1988 Record, using as an example the buck converter. In order to obtain lossless zero-voltage switching at constant switching frequency, the active switch Q1 and rectifier diode D1 in a conventional buck converter shown in FIG. 3(a) are replaced with composite, current bidirectional switches S1 and S2 (realized in practice with MOSFET transistors) as shown in FIG. 3(b). Capacitors across both switches are included in order to model either device parasitic capacitance or externally added capacitor.
A soft-switching buck converter is obtained by using the composite switches S1 and S2 in the basic buck converter topology because lossless soft-transition (zero-voltage switching) in switches S1 and S2 occurs during the time intervals when both switches are turned off, and charge between their capacitors is exchanged in the resonant fashion, ideally without loss. By discharging capacitance across each switch just prior to it being turned on, lossless turn-on switching is provided. While the soft transition from the top switch S1 to the bottom switch S2 is inherently provided by the positive inductor current, transition from the bottom switch S2 to the top switch S1 requires a negative current source to oppose the positive load current flowing through the bottom switch S2. The simplest solution is to design the already existing output inductor L.sub.o such that its current is bidirectional with peak-to-peak magnitude greater than twice the dc load current for all operating conditions of interest as described in Henze, Martin and Parsley, supra. Inductor current waveform during the switching period shown in FIG. 4(a) and equivalent circuits of the converter during two different transition intervals shown in FIGS. 4(b) and 4(c) explain the soft-switching mechanism.
Even though the lossless switching can be achieved in this very simple manner, and voltage stresses on the switches are the same as in the basic PWM converter, the magnitude of the output inductor ripple current, I.sub.L, has to be at least three to four times greater than the maximum load current in order to achieve soft-switching for all operating conditions, particularly at high switching frequency [Henze, Martin and Parsley, supra]. Practical application of such a converter is very limited due to serious drawbacks such as (a) increased conduction losses, (b) high core losses in the output inductor, (c) need for excessive additional output voltage filtering, particularly at high current levels, (d) limited range of the soft-switching frequency, particularly at higher switching frequencies (above 100 kHz), and (e) very low efficiency at light or no-load conditions. Moreover, by operating the converter without soft switching under certain input voltage and load conditions, the size of the EMI filter will be almost the same as in the basic PWM converter.
Another prior-art method, which achieves zero-voltage switching of switches on the primary side of a transformer operated with a constant switching frequency, is a phase-shifted, PWM, full-bridge converter described in R. A. Fisher, K. D. T. Ngo and M. II. Kuo, "A 500 kHz, 250 W Dc-Dc Converter with Multiple Outputs Controlled by Phase-Shifted PWM and Magnetic Amplifiers," High Frequency Power Conversion Conference, pp. 100-110, 1988 Record. Zero-voltage switching was made possible by using phase-shifted (four-state) PWM control, as opposed to the traditional (three-state) PWM control. The converter schematic and ideal waveforms are shown in FIG. 5 and FIG. 6, respectively. Primary side switches S1-S4 are composite switches realized in practice with MOSFET transistors. The inductance L.sub.1 represented either leakage inductance of the transformer or externally added inductance required for one transition interval from passive to active state. Capacitances C.sub.s1 -C.sub.s4 represent parasitic capacitances of the composite switches S1-S4, respectively.
Two switches in the same leg of the bridge (S1, S2 and S3, S4) are driven out of phase at 50% duty ratio with small dead-time t.sub.a and t.sub.b, respectively, (FIG. 6). Their diagonally opposite switches, S4 for S1 and S3 for S2, are driven with delay (or phase shift .theta.) in respect to the corresponding switches S1 and S2 instead of in phase as in the conventional full-bridge converter. Output voltage is then regulated by varying the phase shift .theta. in drive signals of the primary side switches which results in pulse-width modulated voltage applied across the power transformer.
Two transitions, t.sub.a, and t.sub.b, which show ideal waveforms in a phase-shifted, full-bridge converter, are different in nature and duration and correspond to transition of the switches in the left leg (S1 and S2) and the right leg (S3 and S4) of the bridge, respectively. Zero-voltage switching (ZVS) is natural only during transition from active to passive state. Energy for charge displacement of the left leg switch capacitances, C.sub.s1 and C.sub.s2, is provided from the output inductor reflected to the transformer's primary, thus by the load current. On the other hand, zero-voltage switching transition from passive to active state, the right leg transition t.sub.b, is not supported by the load current since the primary current is zero due to simultaneous conduction of the rectifier diodes D1 and D2. In order to provide soft-switching (ZVS) of the right leg switches, S3 and S4, it is necessary to have the inductance in series with the transformer's primary (inductance L.sub.1 in FIG. 5). Energy needed for charge displacement of the right leg switch capacitances, C.sub.s3 and C.sub.s4, is stored in the series (leakage) inductance, L.sub.1, during the active state when the output inductor current is reflected to the primary. In practice, the primary current decays exponentially during passive state (FIG. 6) due to resistance in its circulating path (transistors ON resistances and winding resistances), which further reduces available energy for the left leg transition.
The main drawbacks of the soft-switching, full-bridge converter are (a) there is no soft switching of the rectifier diodes, (b) limited range of zero-voltage switching of the primary side switches, (c) need for feedback isolation and (d) need for high leakage inductance. High leakage inductance has several disadvantages (a) it reduces overall converter efficiency since the leakage field in the transformer produces high eddy current losses in the windings, (b) it limits rate of the current change, di/dt, which results in reduction of the effective duty ratio D.sub.eff and consequently (c) increases primary conduction losses and voltage stress on the output rectifiers due to lower turns ratio required to compensate reduction in effective duty ratio.
Another prior-art method for increasing power density of dc-to-dc converters, particularly in applications where more than one output voltage is required, is to use a single power stage with multiple windings on the power transformer, one for each output voltage. In such a converter, all outputs share the same inverter stage, and only one output, called the main output, is fully regulated by pulse-width modulation of the inverter switches on the primary side. Such a solution then requires feedback isolation, and additional post-regulators are required for independent regulation of the auxiliary outputs against load variation.
One prior-art method uses magnetically controlled saturable reactors, commonly called magamps, as post-regulators due to some benefits as compared to other post-regulation techniques, namely (a) lower parts count, (b) more rugged, (c) more efficient, (d) less EMI noise and (e) lower current stress on the main inverter power switches.
A widely used topology which utilizes magamp post-regulators is a forward converter which requires only one magamp per output. A saturable reactor in a forward converter is reset during the flyback interval of the power transformer. Most of the magnetizing current remains in the primary during flyback of the transformer, and only a small portion is diverted to the secondary for reset of the saturable reactor. In practice, usually up to four magamps can be put in a typical forward converter with no adverse effect upon reset of the power transformer.
In symmetrical topologies (half-bridge, full-bridge and push-pull) unlike the forward converter, two magamps are required per output and the transformer's magnetizing current is not used for reset of the saturable reactors in post-regulators. In a half-bridge converter there is no flyback interval of the transformer, and each saturable reactor is reset during alternate pulses. The magnetizing current shifts to the secondary where it causes a series of problems which make a conventional half-bridge magamp converter less versatile than the forward converter. As a consequence of that, almost all half-bridge and other symmetrical topology with magamp post-regulators require a freewheeling diode, D.sub.F, as shown in FIG. 7.
Without the freewheeling diode D.sub.F, the currents in the main output secondary windings, I.sub.1 and I.sub.2, can be severely unbalanced right after the main switch is turned off. This is a consequence of the simultaneous conduction of both rectifiers, D1 and D2, which short the transformer during freewheeling interval. The situation is different when the main output is lightly loaded. Output inductor L.sub.2 actually supplies power to both the main output and its own load and to the primary of the converter. Therefore, in any practical design of the conventional half-bridge converter (or any other symmetrical topology), the freewheeling diode D.sub.F is essential in the secondary with a magamp post regulator for proper mode of operation.
The freewheeling diode D.sub.F will keep only load current I.sub.02 of the total imbalance from being shifted to the main output, while the imbalance due to magnetizing current of the transformer still remains. In order to maintain continuous current in the main output inductor, which is required for preventing the auxiliary output from sag, it is necessary to provide a path for the magnetizing current. This can be done by (a) increasing minimum load on the main output or (b) providing a shunt path for the excess current. Both approaches reduce converter efficiency and increase converter complexity particularly when auxiliary output is used for supplying disk-drives and fans in computers.
Another undesirable effect is related to the energy stored in the magamp's saturated inductance which is damped back into the transformer at the end of each pulse. If this current, translated by the turns ratio, is greater than the main output's inductor current, the extra pulse will cause the main control loop to shrink the pulse width. As a consequence, the magamp's input pulse may then be inadequate and output voltage will drop.
Conventional half-bridge converter and idealized waveforms are shown in FIG. 8(a). Two equal capacitors C1 and C2 are connected in series across the DC power supply V.sub.g to enable an artificial mid-point P.sub.M to be created. Primary side switches S1 and S2 are driven alternatively during each switching period with duty ratio D=.tau./T.sub.s. Full-wave rectification on the secondary side is provided by center-tapped secondary winding and rectifier diodes D1 and D2. The output voltage V.sub.o is regulated by varying duty ratio D. The transformer's turns ratio is assumed to be n=1.
One can distinguish two successive operating states, active and passive during each half of the switching period T.sub.s (FIG. 8(b)). During the active state .tau. the corresponding pair of the primary switch and rectifier diode (S1-D1 or S2-D2) conduct simultaneously so the primary voltage and current have the same polarity and the power is delivered from the source to the load. Positive voltage of magnitude V.sub.g /2 is applied at the point A and energy is stored in the output inductor L.sub.o. Contrary, during the passive state t.sub.d, both primary side switches are OFF but the rectifier diodes (D1 and D2) conduct inductor current simultaneously and short the transformer. As a consequence of that both primary voltage and current are zero and no power is delivered from the source to the load. Energy stored in the output inductor during active state is now released into the load through the rectifier diodes. This operating state is also called freewheeling state due to the nature of energy transfer.
Soft-switching is not possible because the transformer is shorted by the rectifier diodes just after the one of the primary side switches (S1 or S2) is turned-off so both switches, S1 and S2 are connected instantaneously to the mid-point P.sub.M at voltage V.sub.g /2 and stay there during dead-time t.sub.d. In practice, the switches are implemented with bipolar transistors which have parasitic capacitance represented in FIG. 8(a) by capacitors in parallel with the switches. Thus, capacitors across the primary switches, charged at the V.sub.g /2 during t.sub.d, are therefore, discharged through the switches in lossy manner. It is therefore, necessary to prevent shorting of the transformer during freewheeling states. A search is therefore expected for the topologies which can provide soft-switching by using dc output inductor current with small or even no ripple current, instead of using high ripple, bipolar inductor current as in FIG. 4(a). Good candidates are transformer coupled converter topologies with full-wave rectification on their outputs, which naturally provide alternating current polarity to the primary switches before the beginning of transition intervals. In the following sections we describe the half-bridge converter which is the closest to the soft-switching buck converter from FIG. 3(b), but the same applies for the full-bridge and push-pull converters.